Phase shifter, impedance matching circuit, multi/demultiplexer, and communication terminal apparatus

ABSTRACT

A phase shifter includes a transformer connected between a first port and a second port and including a first coil and a second coil that is magnetically coupled to the first coil, the transformer including a parasitic inductance component; and an impedance adjustment circuit including a reactance element that suppresses a deviation in impedance due to the parasitic inductance component of the transformer. A coupling coefficient between the first coil and the second coil of the transformer and a value of the reactance element of the impedance adjustment circuit are determined such that a phase-shift amount changes in accordance with a frequency band.

CROSS REFERENCE TO RELATED APPLICATIONS

This application claims the benefit of priority to Japanese PatentApplication No. 2015-215081 filed on Oct. 30, 2015, Japanese PatentApplication No. 2015-157569 filed on Aug. 7, 2015, and Japanese PatentApplication No. 2015-062274 filed on Mar. 25, 2015, and is aContinuation Application of PCT Application No. PCT/JP2016/057914 filedon Mar. 14, 2016. The entire contents of each application are herebyincorporated herein by reference.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to a phase shifter provided in ahigh-frequency circuit, and particularly to a phase shifter that shiftsa phase in accordance with a frequency band. The present invention alsorelates to an impedance matching circuit, and particularly to animpedance matching circuit including a phase shifter. The presentinvention also relates to a multi/demultiplexer, and particularly to amulti/demultiplexer including a phase shifter. The present inventionalso relates to a communication terminal apparatus including the phaseshifter, the impedance matching circuit, or the multi/demultiplexer.

2. Description of the Related Art

In general, in a high-frequency circuit, a phase shifter may be used forimpedance matching. In the related art, the phase shifter may be of ahigh-pass-filter type or a low-pass-filter type, and a circuit constantis determined to obtain a desired phase-shift amount at a desiredfrequency. For example, a phase shifter including ahigh-pass-filter-type circuit and a low-pass-filter-type circuit isdisclosed in Japanese Unexamined Patent Application Publication No.2013-98744.

In communication terminal apparatuses and the like typified by a mobilephone terminal, for example, impedance matching is often necessary in aplurality of frequency bands. Considering the case where, as illustratedin FIG. 34, a phase shifter 73 is provided between an impedance matchingcircuit 72 and a second high-frequency circuit 74 and where impedancematching is performed between a first high-frequency circuit 71 and thesecond high-frequency circuit 74 by using the phase shifter 73 and theimpedance matching circuit 72, for example, the phase shifter isrequired to have a phase-shift characteristic in accordance with afrequency band in order to perform impedance matching in a plurality offrequency bands.

For example, if impedance matching is performed in both the low band andthe high band, it may be necessary to largely shift the phase of one ofthe bands while the phase of the other band is almost not shifted. Thereare two phase-shift operations as follows, for example.

(1) The phase of a high-band signal is shifted while the phase of alow-band signal is not shifted.

For example, the pass phase of the low-band signal is about 0° (or180°), and the pass phase of the high-band signal is about 90°.

(2) The phase of a low-band signal is shifted while the phase of ahigh-band signal is not shifted.

For example, the pass phase of the low-band signal is about 90°, and thepass phase of the high-band signal is about 0° or 180°.

Note that in FIG. 34, since the reflected wave from the secondhigh-frequency circuit 74 makes a round trip in the phase shifter 73when seen from the impedance matching circuit 72, the phase-shift amountof the reflected signal in the phase shifter 73 doubles. That is, aphase-shift amount of 90° is necessary in order to obtain a reflectedphase of 180°, and a phase-shift amount of 0° or 180° is necessary inorder to obtain a reflected phase of 0°.

However, as follows, the related art has not provided a phase shifterthat enables phase-shift operations for each frequency band described inthe above (1) and (2).

For example, FIG. 31 illustrates an example of a phase-frequencycharacteristic of a high-pass-filter-type phase shifter illustrated inFIG. 30A. In this example, the phase-shift amount in a low band (700 MHzband) can be 90°, but the phase-shift amount in a high band (2 GHz band)is 30°, not 0°. In addition, FIG. 32 illustrates an example of aphase-frequency characteristic of a low-pass-filter-type phase shifterillustrated in FIG. 30B. In this example, the phase-shift amount in alow band (700 MHz band) can be −90°, but the phase-shift amount in ahigh band (2 GHz band) is about 100°, not 180°. Furthermore, in both thelow band and the high band, the change in the phase-shift amount in thefrequency band is large.

FIG. 33 illustrates an example of an insertion-loss characteristic withrespect to the phase of the high-pass-filter-type phase shifterillustrated in FIG. 30A. Since the phase-shift amount is 180° around thecut-off frequency, the insertion loss is increased if the phase-shiftamount is increased. In addition, in the low-pass-filter-type phaseshifter illustrated in FIG. 30B, if a phase-shift amount of about 180°is obtained in a low band, the cut-off frequency is decreased, and theinsertion loss in a high band is excessively increased.

As described above, in the filter-type phase shifters of the relatedart, it has not been possible to perform phase-shift operations for eachfrequency band described in the above (1) and (2).

On the other hand, in a diplexer or multiplexer provided with filtershaving different frequency characteristics between a common port and aplurality of individual ports, each filter typically cannot obtain anindependent characteristic.

For example, FIG. 36 is a circuit diagram of a diplexer including ahigh-pass filter HPF and a low-pass filter LPF. In this example, anantenna ANT is connected to a common port of the high-pass filter HPFand the low-pass filter LPF. An individual port of the high-pass filterHPF is connected to a high-band circuit, and an individual port of thelow-pass filter LPF is connected to a low-band circuit. The high-passfilter HPF is formed of inductors L11 and L12 connected in shunt to theground and a series-connected capacitor C11, and the low-pass filter LPFis formed of series-connected inductors L21 and L22 and ashunt-connected capacitor C21.

However, in such a circuit illustrated in FIG. 36, when the impedance ofthe inductor L11 of the high-pass filter HPF is excessively decreased ina low frequency band, the inductor L11 substantially becomes a shortcircuit element. Thus, the short circuit element (L11) is connected tothe low-pass filter LPF, and isolation between the low-band circuit andthe high-band circuit is degraded in the low frequency band.

The above-described problem, which arises in a configuration in which ashort circuit substantially occurs in a filter in other frequency bandsamong certain use frequency bands is connected to a common port, arisessimilarly in not only a diplexer formed of the combination of thehigh-pass filter and the low-pass filter but also a multiplexer formedof the combination of a plurality of band-pass filters and the like.

SUMMARY OF THE INVENTION

Preferred embodiments of the present invention provide a phase shiftersuitable for impedance matching, an impedance matching circuit includingthe phase shifter, a multi/demultiplexer in which interference between aplurality of filters is significantly reduced or prevented, and acommunication terminal apparatus including the phase shifter or themulti/demultiplexer.

A phase shifter according to a preferred embodiment of the presentinvention includes a transformer connected between a first port and asecond port and including a first coil and a second coil that ismagnetically coupled to the first coil, the transformer including aparasitic inductance component, and an impedance adjustment circuitincluding a reactance element that suppresses a deviation in impedancedue to the parasitic inductance component of the transformer, wherein acoupling coefficient between the first coil and the second coil of thetransformer and a value of the reactance element of the impedanceadjustment circuit are determined such that a phase-shift amount changesin accordance with a frequency band.

With the above-described configuration combined with an impedancematching circuit, for example, it becomes easy to perform impedancematching in accordance with the frequency band.

A phase shifter according to a preferred embodiment of the presentinvention includes a transformer connected between a first port and asecond port and including a first coil and a second coil that ismagnetically coupled to the first coil, the transformer including aparasitic inductance component, and an impedance adjustment circuitincluding a reactance element that suppresses a deviation in impedancedue to the parasitic inductance component of the transformer, wherein acoupling coefficient between the first coil and the second coil of thetransformer and a value of the reactance element of the impedanceadjustment circuit are determined such that a phase-shift amount in alow band is greater than a phase-shift amount in a high band, that thephase-shift amount in the low band is closer to 180° than 90°, and thatthe phase-shift amount in the high band is closer to 90° than 180°.

By decreasing the difference in impedance between the low band and thehigh band by using the above-described configuration, it becomes easy toperform impedance matching in accordance with the frequency band.

A phase shifter according to a preferred embodiment of the presentinvention includes a transformer connected between a first port and asecond port and including a first coil and a second coil that ismagnetically coupled to the first coil, the transformer including aparasitic inductance component, and an impedance adjustment circuitincluding a reactance element that suppresses a deviation in impedancedue to the parasitic inductance component of the transformer, wherein acoupling coefficient between the first coil and the second coil of thetransformer and a value of the reactance element of the impedanceadjustment circuit are determined such that a phase-shift amount in alow band is greater than a phase-shift amount in a high band, that thephase-shift amount in the low band is closer to 90° than 0°, and thatthe phase-shift amount in the high band is closer to 0° than 90°.

With the above-described configuration combined with an impedancematching circuit including a transformer, it becomes easy to preformimpedance matching in accordance with the frequency band.

It is preferable that the impedance adjustment circuit include a firstcapacitance element connected between the first port connected to thetransformer and a ground, a second capacitance element connected betweenthe second port connected to the transformer and the ground, and a thirdcapacitance element connected between the first port and the second portconnected to the transformer.

With the above-described configuration, although the impedance of thetransformer becomes different from a predetermined value (i.e., about50Ω) in the presence of a parallel parasitic inductance component and aseries parasitic inductance component contained in the transformer, byincluding the reactance element (capacitance elements), the impedance isable to be adjusted.

It is preferable that the impedance adjustment circuit include a firstcapacitance element connected between the first port connected to thetransformer and a ground, a second capacitance element connected betweenthe second port connected to the transformer and the ground, and aseries circuit including a third capacitance element and an inductanceelement, the series circuit being connected between the first port andthe second port connected to the transformer.

With the above-described configuration, it becomes possible for theseries circuit including the third capacitance element and theinductance element to have a predetermined frequency characteristic ofthe phase-shift amount to obtain a predetermined phase-shift amount inaccordance with the frequency in a wide frequency band. In addition,although the impedance of the transformer becomes different from thepredetermined value (i.e., about 50Ω) in the presence of the parallelparasitic inductance component and the series parasitic inductancecomponent contained in the transformer, by including the firstcapacitance element, the second capacitance element, the thirdcapacitance element, and the inductance element, the impedance is ableto be adjusted.

It is preferable that the third capacitance element be mainly defined bya capacitance between the first coil and the second coil. Accordingly, apattern for the formation of the third capacitance element or the thirdcapacitance element as a component is unnecessary, thus reducing thesize and cost.

Is preferable that the first capacitance element be mainly defined by acapacitance between wires of the first coil, and that the secondcapacitance element be mainly defined by a capacitance between wires ofthe second coil. Accordingly, a pattern for the formation of the firstcapacitance element and the second capacitance element or the firstcapacitance element and the second capacitance element as components areunnecessary, thus reducing the size and cost.

It is preferable that a transformer ratio between the first coil and thesecond coil be 1:n (where n is a value other than 1), and that aphase-shift amount of the phase shifter be moved toward a center of aSmith chart by a reflection coefficient (impedance) being moved from ahigh-impedance side to a low-impedance side on the Smith chart andimpedance conversion by the phase shifter.

With the above-described configuration, it becomes possible to shift thephase and to perform impedance conversion by using the transformer, andit becomes possible to have a function of an impedance matching circuitbetween a circuit connected to the first port and a circuit connected tothe second port.

It is preferable that the transformer be provided in a single stack inwhich a plurality of base layers are stacked and that the first coil andthe second coil be defined by conductor patterns provided on the baselayers. Accordingly, it becomes easy to mount a phase shifter as asingle component on a communication terminal apparatus or the likebecause the phase shifter may be mounted on a printed wiring board orthe like.

It is preferable that the first coil and the second coil have the sameor substantially the same inside diameter and the same or substantiallythe same outside diameter and have co-axial coil winding axes.Accordingly, although the first coil and the second coil have a smallnumber of turns, that is, although the first coil and the second coilare small, a transformer with an appropriate coupling coefficient isable to be obtained.

It is preferable that a high-pass filter or a low-pass filter connectedin series to the phase shifter be further included. Accordingly, itbecomes possible to determine a phase-shift amount that cannot beobtained by using only the phase shifter.

It is preferable that the high-pass filter or the low-pass filterinclude a capacitance element and an inductance element, the inductanceelement being magnetically coupled to the first coil or the second coil.With this configuration, it becomes possible to control the frequencycharacteristic of the phase-shift amount.

An impedance matching circuit according to a preferred embodiment of thepresent invention includes the phase shifter according to any one of thepreferred embodiments of the present invention described above, and animpedance matching circuit that is connected in series to the phaseshifter, wherein the impedance matching circuit is a circuit thatprovides impedance matching of an impedance whose phase has been shiftedby the phase shifter.

It is preferable that the phase shifter move the impedance in a low bandto a second quadrant or a third quadrant on a Smith chart, and that theimpedance matching circuit move both impedance in a high band and theimpedance in the low band toward a center of the Smith chart.

With any of the configurations of the impedance matching circuitdescribed above, it becomes easy to perform impedance matching inaccordance with the frequency band.

A multi/demultiplexer according to a preferred embodiment of the presentinvention includes a phase shifter according to any one of theabove-described preferred embodiments of the present invention, ahigh-pass filter that performs high-band passing, and a low-pass filterthat performs low-band passing, wherein the high-pass filter includes ashunt-connected first inductor between a signal line and a ground and afirst capacitor connected in series in a following stage of the firstinductor, the low-pass filter includes a second inductor connected inseries to a common port and a second capacitor connected in shunt to theground in a following stage of the second inductor, the phase shifter isbetween the common port and the first inductor, and the phase shiftershifts a phase such that the high-pass filter is substantially(equivalently) open in a pass frequency band of the low-pass filter seenfrom the common port.

With the above-described configuration, isolation between ports in a lowband is maintained without being affected by the high-pass filter in theuse frequency band (low frequency band) of the low-pass filter.

A multi/demultiplexer according to a preferred embodiment of the presentinvention includes a phase shifter according to any one of theabove-described preferred embodiments of the present invention, and aplurality of SAW (surface acoustic wave) filters including a first SAWfilter and a second SAW filter with mutually different pass frequencybands, the first SAW filter and the second SAW filter each including afirst port and a second port, wherein the first port of the first SAWfilter is connected to a common port via the phase shifter, and thesecond port of the first SAW filter is connected to an individual port,and the phase shifter shifts a phase such that the first SAW filter issubstantially (equivalently) open in a pass frequency band of the secondSAW filter seen from the common port.

With the above-described configuration, isolation between the first SAWfilter and the second SAW filter is maintained in the use frequency bandof the second SAW filter without being affected by the first SAW filterin the use frequency band of the second SAW filter.

A communication terminal apparatus according to a preferred embodimentof the present invention includes a feeder circuit and an antennaconnected to the feeder circuit, wherein, between the feeder circuit andthe antenna, a phase shifter according to any one of the above-describedpreferred embodiments of the present invention, an impedance matchingcircuit according to any one of the above-described preferredembodiments of the present invention, or a multi/demultiplexer accordingto any one of the above-described preferred embodiments of the presentinvention is provided. Accordingly, a communication terminal apparatusin which impedance matching is performed between the antenna element andthe feeder circuit for each predetermined frequency band is obtained. Inaddition, it becomes possible to multiplex and demultiplex signals witha plurality of frequency bands while maintaining isolation betweenports.

According to various preferred embodiments of the present invention, aphase shifter in which the phase-shift amount in accordance with thefrequency band is determined is obtained. In addition, an impedancematching circuit with which impedance matching is easily performed foreach frequency band is obtained. In addition, a multi/demultiplexer inwhich interference between a plurality of filters is significantlyreduced or prevented is obtained. In addition, a communication terminalapparatus in which impedance matching is performed between an antennaelement and a feeder circuit for each predetermined frequency band isobtained. Furthermore, a communication terminal apparatus including amulti/demultiplexer in which isolation between ports is maintained isobtained.

The above and other elements, features, steps, characteristics andadvantages of the present invention will become more apparent from thefollowing detailed description of the preferred embodiments withreference to the attached drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a circuit diagram of a phase shifter 11 according to a firstpreferred embodiment of the present invention.

FIGS. 2A and 2B are each an equivalent circuit diagram of a transformerT.

FIG. 3 illustrates a configuration of an antenna circuit including thephase shifter 11 and an antenna 1.

FIG. 4 illustrates a phase-shift-amount-frequency characteristic of thephase shifter 11 according to the first preferred embodiment of thepresent invention.

FIGS. 5A and 5B illustrate a phase-shift function of the phase shifter11 illustrated in FIG. 3.

FIG. 6A illustrates a locus of an impedance Zm in FIG. 3 in a low bandseen from Pm, and FIG. 6B illustrates a locus of the impedance Zm in ahigh band.

FIG. 7 illustrates a frequency characteristic of a return loss seen fromPm in FIG. 3.

FIG. 8 is an external appearance perspective view of the phase shifter11.

FIG. 9 is a plan view of each layer in the phase shifter 11.

FIG. 10 is a cross-sectional view of the phase shifter 11.

FIG. 11A is a circuit diagram of the phase shifter 11 according to asecond preferred embodiment of the present invention. FIG. 11B is anequivalent circuit diagram illustrating the phase shifter 11 in whichthe ideal transformer IT and parasitic inductance components areseparated from each other.

FIG. 12 illustrates a configuration of an antenna circuit including aphase shifter 13 and the antenna 1 according to a third preferredembodiment of the present invention.

FIG. 13 illustrates a phase-shift-amount-frequency characteristic of thephase shifter 13 according to the third preferred embodiment of thepresent invention.

FIGS. 14A and 14B illustrate a phase-shift function of the phase shifter13 having the characteristic illustrated in FIG. 13.

FIG. 15 is a plan view of each layer in the phase shifter according tothe third preferred embodiment of the present invention.

FIG. 16 is a cross-sectional view of the phase shifter 13.

FIGS. 17A, 17B, and 17C are circuit diagrams of three phase shiftersaccording to a fourth preferred embodiment of the present invention.

FIG. 18 is a circuit diagram of another phase shifter according to apreferred embodiment of the present invention.

FIG. 19 is a circuit diagram of a phase shifter 15 according to a fifthpreferred embodiment of the present invention.

FIG. 20 is a circuit diagram of a phase shifter 16 according to a sixthpreferred embodiment of the present invention.

FIG. 21 is a block diagram of a communication terminal apparatus 200according to a seventh preferred embodiment of the present invention.

FIG. 22 is a circuit diagram of a phase shifter 18 according to aneighth preferred embodiment of the present invention.

FIG. 23 illustrates a frequency characteristic of a phase-shift amountof the phase shifter 18.

FIG. 24A is a circuit diagram illustrating a configuration of a diplexer109 according to a ninth preferred embodiment of the present invention.FIG. 24B is a circuit diagram of a diplexer 109P as a comparativeexample of the diplexer 109.

FIG. 25 illustrates an insertion-loss-frequency characteristic betweenports Pr2 and Pc depending on the presence and absence of an inductorL11 in the diplexer 109P according to the comparative example.

FIG. 26A illustrates, on a Smith chart, areflection-coefficient-frequency characteristic at a predetermined portof the diplexer 109 according to a preferred embodiment of the presentinvention. FIG. 26B illustrates, on a Smith chart, areflection-coefficient-frequency characteristic at a predetermined portof the diplexer 109P according to the comparative example where theinductor L11 is absent. FIG. 26C illustrates, on a Smith chart, areflection-coefficient-frequency characteristic at a predetermined portof the diplexer 109P according to the comparative example where theinductor L11 is present.

FIG. 27A illustrates an insertion-loss-frequency characteristic betweena common port Pc and each of individual ports Pr1 and Pr2 in thediplexer 109 according to a preferred embodiment of the presentinvention. FIG. 27B illustrates an insertion-loss-frequencycharacteristic between the common port Pc and each of the individualports Pr1 and Pr2 in the diplexer 109P according to the comparativeexample where the inductor L11 is absent. FIG. 27C illustrates aninsertion-loss-frequency characteristic between the common port Pc andeach of the individual ports Pr1 and Pr2 in the diplexer 109P accordingto the comparative example where the inductor L11 is present.

FIG. 28 is a circuit diagram illustrating a configuration of amultiplexer 110 according to a tenth preferred embodiment of the presentinvention.

FIG. 29 illustrates, on a Smith chart, areflection-coefficient-frequency characteristic seen from a port of atypical SAW filter.

FIG. 30A is a circuit diagram of a high-pass-filter-type phase shifter,and FIG. 30B is a circuit diagram of a low-pass-filter-type phaseshifter.

FIG. 31 illustrates an example of a phase-frequency characteristic of ahigh-pass-filter-type phase shifter illustrated in FIG. 30A.

FIG. 32 illustrates an example of a phase-frequency characteristic of alow-pass-filter-type phase shifter illustrated in FIG. 30B.

FIG. 33 illustrates an example of an insertion-loss characteristic withrespect to the phase of the high-pass-filter-type phase shifterillustrated in FIG. 30A.

FIG. 34 illustrates an example of a circuit configuration for impedancematching between the first high-frequency circuit 71 and the secondhigh-frequency circuit 74.

FIGS. 35A, 35B, 35C, and 35D each illustrate displacement of animpedance locus in the case where matching is to be performed by amethod of the related art without using the phase shifter.

FIG. 36 is a circuit diagram of a diplexer including a high-pass filterHPF and a low-pass filter LPF.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

Hereinafter, a plurality of preferred embodiments of the presentinvention will be described by taking some specific examples withreference to drawings. In the drawings, like numerals denote likecomponents. In a second preferred embodiment and the following preferredembodiments, elements or features that are common to those in a firstpreferred embodiment will be omitted from description, and differentelements or features will be described. In particular, like functionsand effects obtained by like configurations will not be referred to ineach of the preferred embodiments.

First Preferred Embodiment

FIG. 1 is a circuit diagram of a phase shifter 11 according to the firstpreferred embodiment. The phase shifter 11 includes a transformer T. Thetransformer T includes a first coil L1 and a second coil L2 that ismagnetically coupled to the first coil L1 with a coupling coefficient ofless than 1. The phase shifter 11 further includes an impedanceadjustment circuit including a first capacitance element C1, a secondcapacitance element C2, and a third capacitance element C3.

The first capacitance element C1 is connected in parallel to the firstcoil L1, and the second capacitance element C2 is connected in parallelto the second coil L2. The third capacitance element C3 is connectedbetween the first coil L1 and the second coil L2.

FIGS. 2A and 2B are each an equivalent circuit diagram of thetransformer T. The equivalent circuit of the transformer T can berepresented in several forms. FIG. 2A illustrates an ideal transformerIT, a series parasitic inductance component La that is connected inseries to the upstream side of the ideal transformer IT, a parallelparasitic inductance component Lb that is connected in shunt (parallel)to the upstream side of the ideal transformer IT, and a series parasiticinductance component Lc that is connected in series to the downstreamside of the ideal transformer IT.

FIG. 2B illustrates the ideal transformer IT, two series parasiticinductance components, the series parasitic inductance component La anda series parasitic inductance component Lc1, which are connected inseries to the upstream side of the ideal transformer IT, and theparallel parasitic inductance component Lb that is connected in shunt(parallel) to the upstream side of the ideal transformer IT.

Here, if the transformer ratio of the transformer T is denoted by 1:n,the coupling coefficient between the first coil L1 and the second coilL2 (see FIG. 1) is denoted by k, the inductance of the first coil L1 isdenoted by L1, and the inductance of the second coil L2 is denoted byL2, the inductances of the parasitic inductance components La, Lb, Lc,and Lc1 have the following relationship.

La: L1(1−k)

Lb: k*L1

Lc: L2(1−k)

Lc1: n²*L2*(1−k)

The transformer ratio of the ideal transformer corresponds to atransformer ratio according to the number of turns of the first coil L1and the second coil L2.

In the transformer T according to this preferred embodiment, since thecoupling coefficient k between the first coil L1 and the second coil L2is less than 1, a series inductance component and a parallel inductancecomponent are generated.

FIG. 3 illustrates a configuration of an antenna circuit including thephase shifter 11 and an antenna 1. This antenna circuit includesimpedance matching circuits 41 and 42 and the phase shifter 11 between afeeder circuit 50 and the antenna 1. In FIG. 3, the impedance matchingcircuits 41 and 42 and the phase shifter 11 are examples of an“impedance matching circuit” according to a preferred embodiment of thepresent invention.

In FIG. 3, the phase shifter 11 shifts the phase of a reflected signalfrom the antenna 1 seen from the position denoted by Pa. The impedancematching circuit 41 defines an impedance conversion circuit by using atransformer. For example, the impedance matching circuit 41 increases animpedance Zt seen from the position denoted by Pt to be higher than animpedance Zp seen from the position denoted by Pp toward the antenna 1.The impedance matching circuits 41 and 42 perform impedance matchingbetween the feeder circuit 50 and the antenna 1.

FIG. 4 illustrates a phase-shift-amount-frequency characteristic of thephase shifter 11 according to this preferred embodiment. In thisexample, the phase-shift amount in a low band (700 MHz to 900 MHz band)is substantially 90°, and the phase-shift amount in a high band (1.7 GHzto 2.7 GHz band) is substantially 0°. That is, this preferred embodimentillustrates an example of a phase shifter that does “not shift the phaseof a high-band signal but shifts the phase of a low-band signal”.

FIGS. 5A and 5B illustrate a phase-shift function of the phase shifter11 illustrated in FIG. 3. In FIG. 5A, a locus LBa is the locus of animpedance Za illustrated in FIG. 3 in a low band, and a locus LBp is thelocus of an impedance Zp illustrated in FIG. 3 in a low band. In FIG.5B, a locus HBa is the locus of the impedance Za in a high band, and alocus HBp is the locus of the impedance Zp in a high band.

Since the phase shifter 11 shifts the phase substantially 90° in a lowband as illustrated in FIG. 4, the reflected signal at the positiondenoted by Pp in FIG. 3 is rotated substantially 180° clockwise from thereflected signal at the position denoted by Pa. This corresponds to theimpedance locus illustrated in FIG. 5A being rotated substantially 180°clockwise. Since the phase is almost not shifted in a high band, thereflected signal seen from Pp is the same or substantially the same asthe reflected signal seen from Pa as illustrated in FIG. 5B. In thismanner, in both the low band and the high band, a main portion (majorportion) of the impedance locus is moved to the second quadrant or thethird quadrant on the Smith chart. Here, “the second quadrant on theSmith chart” refers to the upper left region obtained when the Smithchart is divided into four portions in cross-shape, the region in whichthe real part of the reflection coefficient is negative and theimaginary part is positive. In addition, “the third quadrant on theSmith chart” refers to the lower left region obtained when the Smithchart is divided into four portions in cross-shape, the region in whichthe real part of the reflection coefficient is negative and theimaginary part is negative.

The impedance matching circuit 41 illustrated in FIG. 3 is a circuit ofan autotransformer type including a first coil Lp and a second coil Lsthat are magnetically coupled to each other. The impedance matchingcircuit 41 increases impedance seen from the input side at apredetermined impedance conversion ratio. Accordingly, the impedancematching circuit 41 has a function of decreasing the size of the circleof an impedance locus on a Smith chart and shifting the impedance locusclockwise.

FIG. 6A illustrates a locus of an impedance Zm in a low band seen fromPm in FIG. 3, and FIG. 6B illustrates a locus of the impedance Zm in ahigh band. In addition, FIG. 7 illustrates a frequency characteristic ofa return loss seen from Pm in FIG. 3.

In this manner, in both the low band and the high band, the impedance ismoved to the second quadrant or the third quadrant on a Smith chart andthen is moved toward the center of the Smith chart by the impedancematching circuits 41 and 42. Thus, impedance matching is performed inboth the low band and the high band.

The impedance matching circuit 42 changes the impedance mainly in a highband by using a shunt-connected (parallely connected) capacitor or aseries-connected inductor and changes the impedance mainly in a low bandby using a series-connected capacitor and a shunt-connected inductor.

Here, an example of matching by using a shunt-connected inductor or ashunt-connected capacitor without using a phase shifter will bedescribed with reference to FIGS. 35A, 35B, 35C, and 35D.

A locus LBa in FIGS. 35A and 35C is the locus of the impedance Zaillustrated in FIG. 3 in a low band, and a locus HBa in FIGS. 35B and35D is the locus of the impedance Za illustrated in FIG. 3 in a highband. A locus LBb in FIG. 35A is the locus in a low band obtained when ashunt-connected inductor is provided, and a locus HBb in FIG. 35B is thelocus in a low band obtained when a shunt-connected inductor isprovided. The locus LBb in FIG. 35C is the locus in a low band obtainedwhen a shunt-connected capacitor is provided, and the locus HBb in FIG.35D is the locus in a high band obtained when a shunt-connectedcapacitor is provided.

As is clear from FIG. 35A, in the low band where the impedance locus isin the first quadrant on the Smith chart, matching cannot be performedeven if a shunt-connected inductor is provided. In addition, as is clearfrom FIGS. 35A and 35D, in the low band where the impedance locus is inthe first quadrant on the Smith chart, matching is not performed in thehigh band because, although a shunt-connected capacitor is able toperform matching, the shunt-connected capacitor is highly affected inthe high band.

As described above, according to this preferred embodiment, matching isperformed in both the low band and the high band as illustrated in FIGS.6A and 6B.

Second Preferred Embodiment

The second preferred embodiment will describe a specific configurationexample in the phase shifter 11.

FIG. 8 is an external appearance perspective view of the phase shifter11, and FIG. 9 is a plan view of each layer in the phase shifter 11.FIG. 10 is a cross-sectional view of the phase shifter 11. The phaseshifter 11 includes a plurality of insulating bases S1 to S13. Variousconductor patterns are provided on the bases S1 to S13. The phrase“various conductor patterns” refers to, not only a conductor patternprovided on the surface of the base, but also an interlayer connectingconductor. The interlayer connecting conductor includes, not only avia-conductor, but also an end-surface electrode provided on an endsurface of a stack 100.

The top surface of the base S1 corresponds to a surface (bottom surface)on which the stack 100 is mounted. On the base S1, a terminal T1 as afirst port P1, a terminal T2 as a second port P2, a ground terminal GND,and an open terminal NC are provided.

On the bases S7, S6, S5, and S4, conductors L1A1, L1A2, L1A3, and L1A4are provided, respectively. On the base S3, conductors L1A5 and L1B1 areprovided. On the base S2, conductors L1B2 and L1C are provided.

A first end of the conductor L1A1 is connected to the terminal T1defining and functioning as the first port. A second end of theconductor L1A1 is connected to a first end of the conductor L1A2 via avia-conductor V1. A second end of the conductor L1A2 is connected to afirst end of the conductor L1A3 via a via-conductor V2. A second end ofthe conductor L1A3 is connected to a first end of the conductor L1A4 viaa via-conductor V3. A second end of the conductor L1A4 is connected to afirst end of the conductor L1A5 via via-conductor V4. A second end ofthe conductor L1A5 is connected to a first end of the conductor L1B1.The second end of the conductor L1A5 and the first end of the L1B1 areconnected to a first end of the conductor L1B2 via a via-conductor V6. Asecond end of the conductor L1B1 is connected to a second end of theconductor L1B2 via a via-conductor V5. The second end of the conductorL1B2 is connected to a first end of the conductor L1C. A second end ofthe conductor L1C is connected to the ground terminal GND.

On the bases S8, S9, S10, and S11, conductors L2A1, L2A2, L2A3, and L2A4are provided, respectively. On the base S12, conductors L2A5 and L2B1are provided. On the base S13, conductors L2B2 and L2C are provided.

A first end of the conductor L2A1 is connected to the terminal T2defining and functioning as the second port. A second end of theconductor L2A1 is connected to a first end of the conductor L2A2 via avia-conductor V7. A second end of the conductor L2A2 is connected to afirst end of the conductor L2A3 via a via-conductor V8. A second end ofthe conductor L2A3 is connected to a first end of the conductor L2A4 viaa via-conductor V9. A second end of the conductor L2A4 is connected to afirst end of the conductor L2A5 via a via-conductor V10. A second end ofthe conductor L2A5 is connected to a first end of the conductor L2B1.The second end of the conductor L2A5 and the first end of the conductorL2B1 are connected to a first end of the conductor L2B2 via avia-conductor V12. A second end of the conductor L2B1 is connected to asecond end of the conductor L2B2 via a via-conductor V11. The second endof the conductor L2B2 is connected to a first end of the conductor L2C.A second end of the conductor L2C is connected to the ground terminalGND.

The above-described conductors L1A1, L1A2, L1A3, L1A4, L1A5, L1B1, L1B2,and L1C and via-conductors V1, V2, V3, V4, V5, and V6 define the firstcoil L1. The above-described conductors L2A1, L2A2, L2A3, L2A4, L2A5,L2B1, L2B2, and L2C and via-conductors V7, V8, V9, V10, V11, and V12define the second coil L2. Both of the first coil L1 and the second coilL2 preferably are rectangular helical or substantially rectangularhelical coils.

The base layers in the stack 100 may be a non-magnetic ceramic stackformed of low-temperature co-fired ceramics (LTCC) or the like or aresin stack formed of a resin material such as polyimide or a liquidcrystal polymer. The base layers are preferably made of a non-magneticmaterial (not magnetic ferrite) as described above and thus are able tobe applied to a transformer and a phase shifter having a predeterminedinductance and a predetermined coupling coefficient even in ahigh-frequency band over a few hundreds MHz.

The above-described conductor patterns and interlayer connectingconductors are preferably made of a conductor material including Ag orCu as a main component and having a low resistivity. If the base layersare ceramics, for example, the conductor patterns and interlayerconnecting conductors are preferably formed by screen printing andfiring of conductive paste including Ag or Cu as a main component. Ifthe base layers are resin, for example, the conductor patterns andinterlayer connecting conductors are preferably patterned by etching ametal foil such as an Al foil or a Cu foil.

The first coil L1 and the second coil L2 may have the same orsubstantially the same inside diameter and substantially the sameoutside diameter and the same coil winding axis CA (coaxial). However,in this preferred embodiment, a winding axis CA1 of the first coil L1and a winding axis CA2 of the second coil L2 are slightly displacedintentionally. In this preferred embodiment, as illustrated in FIG. 9,the conductors provided on the respective bases define rectangular orsubstantially rectangular loops, for example. However, the conductorsprovided on the bases S5, S4, S3, and S2 have a smaller width in the topand right sides of the loops than in the bottom and left sides.Accordingly, the winding axis CA1 (see FIG. 10) of the first coil L1 isslightly shifted to be upper right of the center of the loop contour. Inaddition, the conductors provided on the bases S10, S11, S12, and S13have a smaller width in the bottom and left sides of the loops than inthe top and right sides. Accordingly, the winding axis CA2 (see FIG. 10)of the second coil L2 is slightly shifted to be lower left of the centerof the loop contour. Thus, the coupling coefficient between the firstcoil L1 and the second coil L2 is intentionally suppressed to be low.

In addition, by providing the bases S6 and S9, the inter-layer distancebetween a main portion of the first coil L1 excluding the conductor L1A1and a main portion of the second coil L2 excluding the conductor L2A1 isincreased. Thus, the coupling coefficient between the first coil L1 andthe second coil L2 is also intentionally suppressed to be low.

FIG. 11A is a circuit diagram of the phase shifter 11 according to apreferred embodiment of the present invention. Here, a transformer isdefined by the first coil L1 and the second coil L2.

The first capacitance element C1 is mainly defined by a capacitancebetween conductor layers provided on the bases S2, S3, S4, S5, S6, andS7. Similarly, the second capacitance element C2 is mainly defined by acapacitance between conductor layers provided on the bases S8, S9, S10,S11, S12, and S13. In addition, the third capacitance element C3 ismainly capacitance between the first coil L1 and the second coil L2,and, in particular, is mainly defined by a capacitance between theconductor L1A1 and the conductor L2A1. In this preferred embodiment, byarranging the conductor L1A1 and the conductor L2A1 to be adjacent toeach other in the stacking direction, the capacitance of the thirdcapacitance element C3 is increased.

The first coil L1 and the second coil L2 are symmetrical in the stackingdirection and have the same number of turns, and accordingly define andfunction as a transformer having an impedance conversion ratio of 1:1.

FIG. 11B is an equivalent circuit diagram illustrating the phase shifter11 in which the ideal transformer IT and parasitic inductance components(series parasitic inductance components La and Lc and parallel parasiticinductance component Lb) are separated from each other.

Although the parasitic inductance components (inductors La, Lb, and Lc)cause the inductance of the transformer to be different from apredetermined value (e.g., about 50Ω), by including the capacitanceelements C1, C2, and C3, the impedance of the transformer is adjusted tobe the predetermined value. In particular, the capacitance elements C1and C2 have a function of correcting a deviation in impedance due to theparallel parasitic inductance component Lb, and the capacitance elementC3 has a function of correcting a deviation in impedance due to theseries parasitic inductance components La and Lc. The above-describedcapacitance elements C1, C2, and C3 are each an example of a “reactanceelement that suppresses a deviation in impedance due to the parasiticinductance component” according to a preferred embodiment of the presentinvention.

As described above, because the coupling coefficient between the firstcoil L1 and the second coil L2 is small, the series parasitic inductancecomponent Lc is large. However, because the capacitance of the thirdcapacitance element C3 is also large, impedance matching is maintained.In addition, because the capacitance of the third capacitance element C3is large, high-band signals pass through the third capacitance elementC3 at a higher proportion than through the transformer defined by thefirst coil L1 and the second coil L2, such that a phase-shift functionof the transformer is almost not performed. This is illustrated also inFIG. 5B in the first preferred embodiment. On the other hand, in alow-band, the amount that bypasses the third capacitance element C3 isrelatively small, such that the phase-shift function of the transformeris performed. Note that the phase-shift amount is less than 180° becausethe coupling coefficient k between the first coil L1 and the second coilL2 is small. The coupling coefficient k is determined to be relativelysmall in order to make the phase-shift amount with respect to a low-bandsignal about 90°.

Note that the positions of the via-conductors V5 and V6 illustrated inFIG. 9 determine the ratio of a parallel connection portion of theconductors L1B1 and L1B2 in the first coil L1 illustrated in FIG. 11A.Similarly, the positions of the via-conductors V11 and V12 illustratedin FIG. 9 determine the ratio of a parallel connection portion of theconductors L2B1 and L2B2 in the second coil L2 illustrated in FIG. 11A.Accordingly, the inductance of the first coil L1 is able to be finelyadjusted depending on the positions of the via-conductors V5 and V6, andthe inductance of the second coil L2 is able to be finely adjusteddepending on the positions of the via-conductors V11 and V12.

Current flows in a dispersed manner in the parallel connection portionof the conductors L1B1 and L1B2, whereas current does not disperse inthis manner in the conductor L1A1. Similarly, current flows in adispersed manner in the parallel connection portion of the conductorsL2B1 and L2B2, whereas current does not disperse in this manner in theconductor L2A1.

Conductor portions of the first coil L1 and the second coil L2, theportions being close to each other in the stacking direction, contributethe most to the coupling. That is, portions of the conductors L1A1 andL2A1, which face each other in the stacking direction over the entirecircumference, contribute to the coupling between the first coil L1 andthe second coil L2. As described above, since current does not dispersein the conductors L1A1 and L2A1 due to the parallel connection portion,the coupling coefficient between the first coil L1 and the second coilL2 is high.

In the above manner, by providing the parallel connection portion apartfrom the conductor pattern of the other coil in the stacking direction,a decrease in the coupling strength caused by the provision of theparallel connection portion is significantly reduced or prevented.

In addition, by arranging the conductors L1A1 and L2A1, which areconnected to the terminals T1 and T2 around the center in the stackingdirection, and by arranging the conductors L1C and L2C, which areconnected to the ground terminal GND, in the upper portion and the lowerportion in the stacking direction, such effects are produced that it ispossible to provide a transformer in which the first coil L1 and thesecond coil L2 share a magnetic flux without a complex configuration andto adjust the capacitance element C3 easily.

Note that, in a case where the coupling coefficient between the firstcoil L1 and the second coil L2 is intentionally decreased, the parallelconnection portion may be provided to be close to the conductor patternof the other coil in the stacking direction in order to use the functionof decreasing the coupling strength by the provision of the parallelconnection portion.

In the first and second preferred embodiments, the coupling coefficientk between the first coil L1 and the second coil L2 is decreased, and thethird capacitance element C3 is increased, such that most of thehigh-band signals pass through the third capacitance element C3. Inaddition, by decreasing the coupling coefficient k between the firstcoil L1 and the second coil L2, the phase-shift amount of thetransformer is significantly reduced or prevented. With theseconfigurations, the phase is shifted 90° in a low band and is almost notshifted in a high band. However, the above configurations are examples.The phase-shift amount in accordance with the frequency band is able tobe determined depending on the coupling coefficient k and thecapacitance of the third capacitance element C3.

In addition, although the first and second preferred embodiments havedescribed an example in which the phase-shift amount in a low band isabout 90° and in which the phase-shift amount in a high band is about0°, as a matter of course, the phase-shift amounts to be determined havea range. If it is determined that the phase-shift amount in a low bandis closer to 90° than 0° and that the phase-shift amount in a high bandis closer to 0° than 90°, the functions and effects described in thefirst and second preferred embodiments are similarly produced.

Third Preferred Embodiment

A third preferred embodiment of the present invention will describe anexample of a phase shifter that does “not shift the phase of a low-bandsignal but shifts the phase of a high-band signal”, unlike the first andsecond preferred embodiments. The circuit diagram of the phase shifteris the same as that illustrated in FIG. 1 in the first preferredembodiment.

FIG. 12 illustrates a configuration of an antenna circuit including aphase shifter 13 and the antenna 1 according to the third preferredembodiment. This antenna circuit includes an impedance matching circuit43 and the phase shifter 13 between the feeder circuit 50 and theantenna 1.

In FIG. 12, the phase shifter 13 shifts the phase of a reflected signalfrom the antenna 1 seen from Zp. The impedance matching circuit 43performs impedance matching between the feeder circuit 50 and theantenna 1 together with the phase shifter 13.

FIG. 13 illustrates a phase-shift-amount-frequency characteristic of thephase shifter 13 according to the third preferred embodiment. In thisexample, the phase-shift amount in a low band (700 MHz to 900 MHz band)is about 180°, and the phase-shift amount in a high band (1.7 GHz to 2.7GHz band) is about 90°.

FIGS. 14A and 14B illustrate a phase-shift function of the phase shifter13 having the characteristic illustrated in FIG. 13. The locus LBa inFIG. 14A is the locus of the impedance Za in a low band seen from Pa inFIG. 12, and the locus LBp is the locus of the impedance Zp in a lowband seen from Pp in FIG. 12. The locus HBa in FIG. 14B is the locus ofthe impedance Za in a high band seen from Pa in FIG. 12, and the locusHBp is the locus of the impedance Zp in a high band seen from Pp in FIG.12.

Although the coupling coefficient k between the first coil L1 and thesecond coil L2 is intentionally decreased in order to decrease thephase-shift amount obtained by the transformer configuration in thefirst and second preferred embodiments, the coupling coefficient kbetween the first coil L1 and the second coil L2 is made to be as closeto 1 as possible in order to make the phase-shift amount obtained by thetransformer configuration close to 180° in the third preferredembodiment. As illustrated in FIG. 13, the phase shifter according tothis preferred embodiment shifts the phase about 180° (about 360° inround trip) in a low band, and accordingly, as illustrated in FIG. 14A,the phase of the reflected signal seen from Pp is substantially the sameas the phase of the reflected signal seen from Pa. As for a high band,the phase of the reflected signal seen from Pp is rotated about 180°clockwise from the phase of the reflected signal seen from Pa. Inaddition, the capacitance component of the second capacitance element C2(see FIGS. 11A and 11B) results in the impedance locus having a smallcircle.

In this manner, because of the phase shifter 13, the impedance in thelow band is almost not moved on the Smith chart, and the impedance inthe frequency band around the center is mainly on a high-impedance side.In addition, the impedance in a high frequency band around the center ismoved to a high-impedance side on the Smith chart, i.e., a positionclose to the impedance in a low band. The impedance matching circuit 43illustrated in FIG. 12 includes a series-connected reactance element anda shunt-connected reactance element and causes the impedance illustratedin FIGS. 14A and 14B to be matched with the impedance of the feedercircuit 50.

In the above manner, by making the impedance in a low frequency bandaround the center substantially the same as and the impedance in a highfrequency band around the center, changes in the impedance in thetransformer and the other circuit elements are able to be made to besubstantially the same, thus making it easier to perform impedancematching.

The external appearance of the phase shifter 13 according to thispreferred embodiment is the same as that illustrated in FIG. 8 in thesecond preferred embodiment.

FIG. 15 is a plan view of each layer in the phase shifter 13 accordingto this preferred embodiment. FIG. 16 is a cross-sectional view of thephase shifter 13. The circuit diagram of the phase shifter 13 is thesame as that in FIG. 11A illustrated in the second preferred embodiment.

The phase shifter 13 includes the plurality of insulating bases S1 toS9. Various conductor patterns are provided on the bases S1 to S9. Thephrase “various conductor patterns” refers to, not only a conductorpattern provided on the surface of the base, but also an interlayerconnecting conductor. The interlayer connecting conductor includes, notonly a via-conductor, but also an end-surface electrode provided on anend surface of a stack.

The top surface of the base S1 corresponds to a surface (bottom surface)on which the stack is mounted. On the base S1, the terminal T1 as thefirst port P1, the terminal T2 as the second port P2, the groundterminal GND, and then open terminal NC are provided.

On the bases S5 and S4, the conductors L1A1 and L1A2 are provided,respectively. On the base S3, the conductors L1A3 and L1B1 are provided.On the base S2, the conductors L1B2 and L1C are provided.

A first end of the conductor L1A1 is connected to the terminal T1defining and functioning as the first port. A second end of theconductor L1A1 is connected to a first end of the conductor L1A2 via thevia-conductor V1. A second end of the conductor L1A2 is connected to afirst end of the conductor L1A3 via the via-conductor V2. A second endof the conductor L1A3 is connected to a first end of the conductor L1B1.The second end of the conductor L1A3 and the first end of the L1B1 areconnected to a first end of the conductor L1B2 via the via-conductor V3.A second end of the conductor L1B1 is connected to a second end of theconductor L1B2 via the via-conductor V4. The second end of the conductorL1B2 is connected to a first end of the conductor L1C. A second end ofthe conductor L1C is connected to the ground terminal GND.

On the bases S6 and S7, the conductors L2A1 and L2A2 are provided,respectively. On the base S8, the conductors L2A3 and L2B1 are provided.On the base S9, the conductors L2B2 and L2C are provided.

A first end of the conductor L2A1 is connected to the terminal T2defining and functioning as the second port. A second end of theconductor L2A1 is connected to a first end of the conductor L2A2 via thevia-conductor V5. A second end of the conductor L2A2 is connected to afirst end of the conductor L2A3 via the via-conductor V6. A second endof the conductor L2A3 is connected to a first end of the conductor L2B1.The second end of the conductor L2A3 and the first end of the conductorL2B1 are connected to a first end of the conductor L2B2 via thevia-conductor V7. A second end of the conductor L2B1 is connected to asecond end of the conductor L2B2 via the via-conductor V8. The secondend of the conductor L2B2 is connected to a first end of the conductorL2C. A second end of the conductor L2C is connected to the groundterminal GND.

The above-described conductors L1A1, L1A2, L1A3, L1B1, L1B2, and L1C andvia-conductors V1, V2, V3, and V4 define the first coil L1. Theabove-described conductors L2A1, L2A2, L2A3, L2B1, L2B2, and L2C andvia-conductors V5, V6, V7, and V8 define the second coil L2. Both of thefirst coil L1 and the second coil L2 preferably are rectangular helicalor substantially rectangular helical coils.

The conductors L1A1 and L2A1 on the bases S5 and S6 have a smaller widththan the other conductors. In addition, the conductors L1A2 and L2A2 onthe bases S4 and S7 have a smaller width than the conductors on thebases S3 and S8. Accordingly, the capacitance between the first coil L1and the second coil L2 is suppressed to be low, and the capacitance ofthe third capacitance element C3 is suppressed to be low. In addition,in this preferred embodiment, among the conductor patterns in theplurality of layers defining the first coil L1 and the conductorpatterns in the plurality of layers defining the second coil L2, thecloser the conductor patterns are, the smaller the width is; and themore distant the conductor patterns are, the larger the width is. Byhaving such a relationship, the first coil L1 and the second coil L2 donot have an excessively small narrow average width, and the capacitancebetween the first coil L1 and the second coil L2 is suppressed. Thus, aconductor loss is reduced, and an increase in the insertion loss isreduced.

The first coil L1 and the second coil L2 have the same or substantiallythe same inside diameter and the same or substantially the same outsidediameter and the same coil winding axis CA (coaxial). In addition,unlike in the phase shifter 11 described in the second preferredembodiment, the inter-layer distance between the layers of the firstcoil L1 and the layers of the second coil L2 is close. Thus, atransformer having a high coupling coefficient k between the first coilL1 and the second coil L2 is obtained.

In addition, although this preferred embodiment has described an examplein which the phase-shift amount in a low band is about 180° and in whichthe phase-shift amount in a high band is about 90°, as a matter ofcourse, the phase-shift amounts to be determined have a range. If it isdetermined that the phase-shift amount in a low band is closer to 180°than 90° and that the phase-shift amount in a high band is closer to 90°than 180°, the functions and effects described in this preferredembodiment are produced.

Fourth Preferred Embodiment

A fourth preferred embodiment of the present invention will describe aphase shifter including a high-pass filter and a low-pass filter.

FIGS. 17A, 17B, and 17C are circuit diagrams of three phase shiftersaccording to the fourth preferred embodiment. In the example illustratedin FIG. 17A, a high-pass filter 61 is connected in series to the phaseshifter 11. The high-pass filter 61 includes a series-connectedcapacitance element C4 and a shunt-connected inductance element L3. Inthe example illustrated in FIG. 17B, a high-pass filter 62 is connectedin series to the phase shifter 11. The high-pass filter 62 includes theseries-connected capacitance element C4 and a parallely connectedinductance element L3 and a parallely connected capacitance element C5.In the example illustrated in FIG. 17C, a low-pass filter 63 isconnected in series to the phase shifter 11. The low-pass filter 63includes a series-connected inductance element L4, a parallely connectedcapacitance element C6, and a parallely connected capacitance elementC7.

By connecting the high-pass filter or the low-pass filter in series tothe phase shifter 11 as in this preferred embodiment, if a predeterminedphase-shift amount is not obtained by using only the phase shifter 11,the lacking phase-shift amount is able to be compensated for by usingthe high-pass filter or the low-pass filter, thus providing the phaseshifter having the predetermined phase-shift amount.

For example, if the phase shifter having a transformer configurationshifts the phase 175° and the phase-shift amount is to be adjusted to180°, the phase-shift amount is increased by 5° by using the high-passfilter that is additionally provided as illustrated in FIG. 17A. Theadded phase-shift amount is so small that an increase in the loss isalmost not generated. Depending on the phase-shift amount to be added, amatching deviation may occur in some cases. In those cases, matching isadjusted by using the shunt-connected capacitance element C5 asillustrated in FIG. 17B. In contrast, if the phase-shift amount is to beincreased, a low-pass filter is additionally provided as illustrated inFIG. 17C.

Note that the above-described inductance elements and capacitanceelements may be individual components or may be defined by conductorpatterns. In addition, the above-described inductance elements andcapacitance elements may be formed integrally with the phase shifter 11.In the phase shifters illustrated in FIGS. 17A, 17B, and 17C, byintegrally providing the inductance elements L3 and L4 in the filterwith the phase shifter 11, the inductance elements L3 and L4 may bemagnetically coupled to the first coil L1 and the second coil L2.Accordingly, in the examples illustrated in FIGS. 17A and 17B, thephase-shift amounts obtained by additionally providing the high-passfilters 61 and 62 may be differentiated from those in the case where theabove coupling is not obtained. Similarly, in the example illustrated inFIG. 17C, the phase-shift-amount-frequency characteristic obtained byadditionally providing the low-pass filter 63 may be differentiated fromthat in the case where the above coupling is not obtained.

FIG. 18 is a circuit diagram of another phase shifter according to thispreferred embodiment. The basic configuration of the circuit is the sameas that of the circuit illustrated in FIG. 17A, but the polarity of theinductance element L3 coupled to the first coil L1 and the second coilL2 is opposite to that in the circuit illustrated in FIG. 17A. With thecoupling having this polarity, an increase and a decrease in thephase-shift amount obtained by additionally providing the high-passfilter 61 is also able to be adjusted. As for the circuits illustratedin FIGS. 17B and 17C, depending on the polarity of the coupledinductance elements L3 and L4, an increase and a decrease in thephase-shift amount obtained by additionally providing the high-passfilter 62 and the low-pass filter 63 is also able to be adjusted.

Fifth Preferred Embodiment

A fifth preferred embodiment of the present invention will describe anexample of a phase shifter that shifts the phase and converts theimpedance.

FIG. 19 is a circuit diagram of a phase shifter 15 according to thefifth preferred embodiment. Although the first preferred embodiment hasdescribed a phase shifter including a transformer with an impedanceconversion ratio of 1:1 in the examples illustrated in FIG. 1 and FIGS.2A and 2B, the impedance conversion ratio may be 1:n (where n is a valueother than 1). For example, if n<1 is satisfied, an antenna havingimpedance lower than the impedance of the feeder circuit is able to bematched with the impedance of the feeder circuit. Therefore, accordingto this preferred embodiment, it is possible to perform a predeterminedphase shift and impedance matching.

Sixth Preferred Embodiment

FIG. 20 is a circuit diagram of a phase shifter 16 according to a sixthpreferred embodiment of the present invention. The phase shifter 16according to this preferred embodiment includes an autotransformerincluding the first coil L1 and the second coil L2 that are magneticallycoupled to each other. The first capacitance element C1 is connectedbetween the first port P1 and the ground, and the second capacitanceelement C2 is connected between the second port P2 and the ground. Inaddition, the third capacitance element C3 is connected between thefirst port P1 and the second port P2.

In the autotransformer such as the transformer in this preferredembodiment, since the coupling coefficient between the first coil L1 andthe second coil L2 is less than 1, a parallel inductance component and aseries inductance component are generated. In addition, the impedancematching is performed by using the capacitance elements C1, C2, and C3.

Seventh Preferred Embodiment

A seventh preferred embodiment of the present invention will describe acommunication terminal apparatus. FIG. 21 is a block diagram of acommunication terminal apparatus 200 according to the seventh preferredembodiment. The communication terminal apparatus 200 according to thispreferred embodiment includes the antenna 1, an antenna matching circuit40, a phase-shift circuit 30, a communication circuit 51, a basebandcircuit 52, an application processor 53, and an input/output circuit 54.The communication circuit 51 includes a transmission circuit and areception circuit for a low band (700 MHz to 1.0 GHz) and a high band(1.4 GHz to 2.7 GHz) and an antenna duplexer, for example. The antenna 1is a monopole antenna, an inverse L antenna, an inverse F antenna, orthe like compatible with the low band and the high band.

The above-described components are preferably stored in a singlehousing. For example, the antenna matching circuit 40, the phase-shiftcircuit 30, the communication circuit 51, the baseband circuit 52, andthe application processor 53 are mounted on a printed wiring board, andthe printed wiring board is stored in the housing. The input/outputcircuit 54 is incorporated in the housing as a display and touch panel.The antenna 1 is mounted on the printed wiring board or arranged on aninner wall of the housing or inside the housing.

The communication terminal apparatus having the above-describedconfiguration and including an antenna that performs matching in a wideband is obtained.

Eighth Preferred Embodiment

An eighth preferred embodiment of the present invention will describe aphase shifter in which the phase-shift amount has a frequencycharacteristic. As described in some of the above-described preferredembodiments, the transformer shifts the phase 180° but does not have aphase-shift-amount-frequency characteristic. Accordingly, it isdifficult to obtain a predetermined phase-shift amount in a specificfrequency band by using only the transformer or to obtain apredetermined phase-shift amount in accordance with the frequency in afrequency range.

FIG. 22 is a circuit diagram of a phase shifter 18 according to theeighth preferred embodiment. The phase shifter 18 is different from thephase shifter 11 illustrated in FIG. 1 in the first preferred embodimentin that an inductance element L5 is provided. That is, a series circuitSR including the third capacitance element C3 and the inductance elementL5 is provided between the first port P1 and the second port P2connected to the transformer T. The other basic configuration is thesame as that of the phase shifter 11 according to the first preferredembodiment.

As illustrated in FIG. 22, by providing the LC series circuit SRincluding the third capacitance element C3 and the inductance element L5in parallel to the transformer T (as a bypass line), the phase shifter18 according to this preferred embodiment includes a low-pass filterportion LPF and a high-pass filter portion HPF. That is, the firstcapacitance element C1, the second capacitance element C2, and theinductance element L5 form the low-pass filter portion LPF, and thefirst coil L1, the second coil L2, and the third capacitance element C3define the high-pass filter portion HPF. Alternatively, a parallelparasitic inductance component (see Lb in FIGS. 2A and 2B) of thetransformer T including the first coil L1 and the second coil L2 and thethird capacitance element C3 is able to define the high-pass filter HPF.

FIG. 23 illustrates a frequency characteristic of each of the phaseshifter 18 according to this preferred embodiment and a phase shifteraccording to a comparative example. In the phase shifter according tothe comparative example, the inductance element L5 in FIG. 22 is notprovided, and the third capacitance C3 is provided in a bypass line.

In FIG. 23, a curve PS(LC) denotes the frequency characteristic of thephase shifter 18, and a curve PSC denotes the frequency characteristicof the phase shifter according to the comparative example. The phaseshifter according to the comparative example functions as atransformer-phase shifter in a low frequency band. In a high frequencyband, a larger amount of signals bypasses the third capacitance elementC3, and the phase-shift amount approaches 0°.

In contrast, in the phase shifter 18 according to this preferredembodiment, the phase-shift amount is negative in high frequencies. FIG.23 illustrates functions of the phase shifter 18 according to thispreferred embodiment in three frequency bands F1, F2, and F3 separatelyas described below.

In the low frequency band F1, the capacitance of the third capacitanceelement C3 is dominant in the LC series circuit SR. Accordingly, signalsthat travel between the ports P1 and P2 almost do not bypass through theLC series circuit SR. That is, the characteristic of the transformer Tappears.

In the middle frequency band F2, the capacitance of the thirdcapacitance element C3 is dominant over the inductance element L5 in theLC series circuit SR, and the LC series circuit SR is capacitive.Accordingly, the bypass circuit defines and functions as a high-passfilter, and the phase-shift amount is decreased with an increase in thefrequency.

In the high frequency band F3, the inductance of the inductance elementL5 is dominant over the third capacitance element C3 in the LC seriescircuit SR, and the LC series circuit SR is inductive. Accordingly, thebypass circuit defines and functions as a low-pass filter, and thephase-shift amount is negative. The frequency at which the phase-shiftamount is 0° corresponds to the series resonance frequency of the LCseries circuit SR.

The above-described phase-shift-amount-frequency characteristic isdetermined depending on the first capacitance element C1, the secondcapacitance element C2, the third capacitance element C3, the inductanceelement L5, and a parallel parasitic inductance component of thetransformer T.

In the above manner, it is possible for the phase-shift amount to have apredetermined wide frequency characteristic. In addition, it is possibleto obtain a predetermined phase-shift amount in accordance with thefrequency in a wide frequency band.

In addition, each of the first capacitance element C1, the secondcapacitance element C2, the third capacitance element C3, and theinductance element L5 does not only determine the frequencycharacteristic of the phase-shift amount, but also has a function of anelement for matching the impedance to be a predetermined impedance(typically about 50Ω).

Ninth Preferred Embodiment

A ninth preferred embodiment of the present invention will describe adiplexer including a high-pass filter, a low-pass filter, and a phaseshifter. The diplexer is an example of a “multi/demultiplexer” accordingto a preferred embodiment of the present invention.

FIG. 24A is a circuit diagram illustrating a configuration of a diplexer109 according to the ninth preferred embodiment. The diplexer 109includes the high-pass filter HPF that performs high-band passing andthe low-pass filter LPF that performs low-band passing connected betweena common port Pc and each of individual ports Pr1 and Pr2. In thispreferred embodiment, the antenna 1 is connected to the common port P.

The high-pass filter HPF includes a first inductor L11 connected inshunt between a signal line and the ground and a first capacitor C11connected in series to the line in the following stage of the firstinductor L11. In this preferred embodiment, the high-pass filter HPFfurther includes an inductor L13 connected in parallel to the firstcapacitor C11 and further includes a shunt-connected inductor L12 in thefollowing stage.

The low-pass filter LPF includes a second inductor L21 connected inseries to the common port Pc and a second capacitor C21 connected inshunt to the ground in the following stage of the second inductor L21.In this preferred embodiment, the low-pass filter LPF further includes acapacitor C22 connected in parallel to the second inductor L21 andfurther includes a series-connected parallel connection circuitincluding an inductor L22 and a capacitor C23 in the following stage ofthe shunt-connected second capacitor C21.

A phase shifter 19 is inserted between the common port Pc and the firstinductor L11. The phase shifter 19 shifts the phase such that thehigh-pass filter HPF is substantially (equivalently) open in a passfrequency band (low band) of the low-pass filter LPF seen from thecommon port Pc.

FIG. 24B is a circuit diagram of a diplexer 109P as a comparativeexample of the diplexer 109. Unlike the diplexer 109, the diplexer 109Pdoes not include the phase shifter 19.

FIG. 25 illustrates an insertion-loss-frequency characteristic betweenthe ports Pr2 and Pc depending on the presence and absence of theinductor L11 in the diplexer 109P according to the comparative example.Here, a characteristic curve C denotes the characteristic obtained inthe presence of the inductor L11, and a characteristic curve N denotesthe characteristic obtained in the absence of the inductor L11.

If the inductor L11 connected in shunt to the ground and connected tothe common port Pc is absent, isolation is high in a low band. However,because of the absence of the inductor L11, the high-pass filter HPF hasa degraded characteristic (which will be described later).

FIG. 26A illustrates, on a Smith chart, areflection-coefficient-frequency characteristic at a predetermined portof the diplexer 109 according to this preferred embodiment. FIG. 26Billustrates, on a Smith chart, a reflection-coefficient-frequencycharacteristic at a predetermined port of the diplexer 109P according tothe comparative example where the inductor L11 is absent. FIG. 26Cillustrates, on a Smith chart, a reflection-coefficient-frequencycharacteristic at a predetermined port of the diplexer 109P according tothe comparative example where the inductor L11 is present.

In each of FIGS. 26A, 26B, and 26C, a curve A denotes the characteristicseen from the common port Pc, and a curve F denotes the characteristicseen from the individual port Pr1. In addition, each marker and thefrequency have the following relationship.

m1, m3: 960 MHz

m2, m4: 1.7 GHz

As is clear from the comparison between FIGS. 26A, 26B, and 26C, in thediplexer 109 according to this preferred embodiment, the filter issubstantially open in a low band (960 MHz) seen from the common port PC,and matching to a predetermined impedance (50Ω) is performed in a highband (1.7 GHz). In addition, matching to the predetermined impedance(50Ω) is performed in a high band (1.7 GHz) seen from the individualport Pr1.

FIG. 27A illustrates an insertion-loss-frequency characteristic betweenthe common port Pc and each of the individual ports Pr1 and Pr2 in thediplexer 109 according to this preferred embodiment. FIG. 27Billustrates an insertion-loss-frequency characteristic between thecommon port Pc and each of the individual ports Pr1 and Pr2 in thediplexer 109P according to the comparative example where the inductorL11 is absent. FIG. 27C illustrates an insertion-loss-frequencycharacteristic between the common port Pc and each of the individualports Pr1 and Pr2 in the diplexer 109P according to the comparativeexample where the inductor L11 is present.

In FIGS. 27A, 27B, and 27C, a curve LPF denotes the characteristic ofthe low-pass filter LPF, a curve HPF denotes the characteristic of thehigh-pass filter HPF, and a curve ISO denotes the characteristic ofisolation between ports.

As is clear from the comparison between FIGS. 27A, 27B, and 27C, in thediplexer 109P according to the comparative example where the inductorL11 is absent, as illustrated in FIG. 27B, the obtained isolation ISObetween ports is only about −10 dB. In the diplexer 109P according tothe comparative example where the inductor L11 is present, asillustrated in FIG. 27C, the insertion loss of the low-pass filter LPFin a low band (greater than or equal to 700 MHz and less than or equalto 960 MHz) reaches −5 dB.

In contrast, in the diplexer 109 according to this preferred embodiment,the insertion loss of the low-pass filter LPF in the low band is lessthan or equal to −1 dB, and the attenuation amount in a high band(greater than or equal to 1.7 GHz and less than or equal to 2.7 GHz) isgreater than or equal to −30 dB. In addition, the insertion loss of thehigh-pass filter HPF in a high band is less than or equal to −1 dB, andthe attenuation amount in a low band is greater than or equal to −28 dB.

Tenth Preferred Embodiment

A tenth preferred embodiment of the present invention will describe amultiplexer including a plurality of SAW filters and a phase shiftertogether with these SAW filters. This multiplexer is an example of the“multi/demultiplexer” according to a preferred embodiment of the presentinvention.

FIG. 28 is a circuit diagram illustrating a configuration of amultiplexer 110 according to the tenth preferred embodiment. Themultiplexer 110 includes phase shifters 19 a, 19 b, 19 c, and 19 d andSAW filters SAWa, SAWb, SAWc, and SAWd, each of which is connectedbetween the common port Pc and each of individual ports Pr1, Pr2, Pr3,and Pr4. Each of the phase shifters 19 a, 19 b, 19 c, and 19 d is thetransformer-type phase shifter described in some of the above preferredembodiments. In this preferred embodiment, for example, an antenna isconnected to the common port Pc, and a communication circuit for acorresponding frequency band is connected to each of the individualports Pr1, Pr2, Pr3, and Pr4.

Each of the SAW filters SAWa, SAWb, SAWc, and SAWd includes a first portand a second port, and the pass frequency bands thereof are differentfrom one another. The first port of the first SAW filter SAWa isconnected to the common port Pc via the phase shifter 19 a, and thesecond port thereof is connected to the individual port Pr1. Similarly,the first port of the second SAW filter SAWb is connected to the commonport Pc via the phase shifter 19 b, and the second port thereof isconnected to the individual port Pr2; the first port of the third SAWfilter SAWc is connected to the common port Pc via the phase shifter 19c, and the second port thereof is connected to the individual port Pr3;and the first port of the fourth SAW filter SAWd is connected to thecommon port Pc via the phase shifter 19 d, and the second port thereofis connected to the individual port Pr4.

For example, the center frequency of the pass band of the first SAWfilter SAWa is 700 MHz, the center frequency of the pass band of thesecond SAW filter SAWb is 800 MHz, and the center frequency of the passband of the third SAW filter SAWc is 900 MHz. In addition, the centerfrequency of the pass band of the fourth SAW filter SAWd is 2 GHz. Thatis, the SAW filters SAWa, SAWb, and SAWc are for a low band, and the SAWfilter SAWd is for a high band.

The phase shifter 19 a shifts the phase such that the first SAW filterSAWa is substantially open in pass frequency bands of the SAW filtersSAWb, SAWc, and SAWd, which are other than the first SAW filter SAWa,seen from the common port Pc. In addition, the phase shifter 19 b shiftsthe phase such that the second SAW filter SAWb is substantially open inpass frequency bands of the SAW filters SAWa, SAWc, and SAWd, which areother than the second SAW filter SAWb, seen from the common port Pc. Thephase shifter 19 c shifts the phase such that the third SAW filter SAWcis substantially open in pass frequency bands of the SAW filters SAWa,SAWb, and SAWd, which are other than the third SAW filter SAWc, seenfrom the common port Pc. Similarly, the phase shifter 19 d shifts thephase such that the fourth SAW filter SAWd is substantially open in passfrequency bands of the SAW filters SAWa, SAWb, and SAWc, which are otherthan the fourth SAW filter SAWd, seen from the common port Pc.

FIG. 29 illustrates, on a Smith chart, areflection-coefficient-frequency characteristic seen from a port of atypical SAW filter. The impedance is substantially short-circuited in afrequency band lower than the pass band, the predetermined impedance(i.e., about 50Ω) is obtained at a center frequency fc of the passfrequency band, and the impedance is substantially short-circuited againin a frequency band higher than the pass band.

Accordingly, if a plurality of SAW filters with markedly different passfrequency bands are directly connected to a common port, in a usefrequency band, the common port Pc is substantially short-circuited tothe ground. Therefore, for example, a low-band SAW filter and ahigh-band SAW filter are regarded as being short-circuited to each otherand thus cannot be directly connected to the common port Pc.

According to this preferred embodiment, even if SAW filters withmarkedly different pass frequency bands are used, since the phaseshifter shifts the phase about 180°, the SAW filters are regarded asbeing open to each other. Therefore, the SAW filters can be directlyconnected to the common port Pc via the phase shifters. In this state,isolation between ports is maintained.

The phase-shift amount of each of the above-described phase shifters 19a to 19 d is not limited to 180°, and an appropriate phase-shift amountin accordance with the frequency band may be determined. For example, itis possible to use a transformer-type phase shifter in which thephase-shift amount has a frequency characteristic, which has beendescribed in the second and eighth preferred embodiments.

Considering the case where each of the phase shifters 19 a to 19 dillustrated in FIG. 28 includes a high-pass-filter-type phase shifter ofthe related art, the frequency characteristic has a steep slope(frequency characteristic is large) due to the characteristics of L andC. Accordingly, a predetermined phase-shift amount can be obtained onlyin a narrow band, and thus it is not possible to maintain isolationbetween ports in a wide band. Therefore, in the related art, if aplurality of SAW filters with markedly different frequency bands areused, a circuit has been configured such that a diplexer demultiplexes ahigh-band signal and a low-band signal and that the plurality of SAWfilters are switched by using a switch for each band.

The mechanism to achieve phase adjustment according to a preferredembodiment of the present invention uses a transformer. Therefore,comparing the mechanism according to a preferred embodiment of thepresent invention with mechanisms for phase adjustment of ahigh-pass-filter type, a low-pass-filter type, and a line type, thechange in the phase-shift amount depending on the change in thefrequency is small. Accordingly, the phase is able to be made to beinverse in a larger band than in the related art in order to open afilter, and it is possible to connect an SAW filter in a larger band.

For example, by designing a configuration such that the phase is shifted180° to be inverse in a band lower than the frequency used by the SAWfilter, that the phase is substantially 0° in a pass band of the SAWfilter, and that the phase is shifted −180° in a high frequency band, itis possible to multiplex and demultiplex signals with markedly differentfrequency bands while maintaining isolation between ports without usinga diplexer or a switch.

Other Preferred Embodiments

Some of the above-described preferred embodiments of the presentinvention have illustrated examples in which the impedance adjustmentcircuit that adjusts the impedance of the transformer includes the threecapacitance elements C1, C2, and C3. The impedance adjustment circuit isa circuit that corrects or actively modifies the displacement in theimpedance due to the parallel inductance component and the seriesinductance component, which are parasitic components of the transformer.Accordingly, the impedance adjustment circuit is not necessarily definedby the three capacitance elements. The impedance of the transformer Tmay be finely adjusted by connecting the transformer to a predeterminedreactance element in parallel or in series.

Note that in the above-described preferred embodiments of the presentinvention, the first capacitance element C1 and the second capacitanceelement C2 are not limited to capacitance between wires of coils, andconductor patterns other than coils may be provided. In addition, acapacitor may be connected as an externally attached component.Furthermore, the third capacitance element C3 is not limited tocapacitance between coils, and conductor patterns other than coils maybe provided. In addition, a capacitor may be connected as an externallyattached component.

Lastly, the above-described preferred embodiments of the presentinvention are illustrative in all points and are not limiting. A personskilled in the art may make changes or modifications as appropriate. Forexample, elements or features of the configurations described indifferent preferred embodiments may be replaced or combined with eachother. The scope of the present invention is to be defined by the scopeof the claims, not the above-described preferred embodiments. Inaddition, the scope of the present invention is to include equivalentsto the scope of the claims and all changes within the scope of theclaims.

While preferred embodiments of the present invention have been describedabove, it is to be understood that variations and modifications will beapparent to those skilled in the art without departing from the scopeand spirit of the present invention. The scope of the present invention,therefore, is to be determined solely by the following claims.

What is claimed is:
 1. A phase shifter comprising: a transformerconnected between a first port and a second port and including a firstcoil and a second coil that is magnetically coupled to the first coil,the transformer including a parasitic inductance component; and animpedance adjustment circuit including a reactance element thatsuppresses a deviation in impedance due to the parasitic inductancecomponent of the transformer; wherein a coupling coefficient between thefirst coil and the second coil of the transformer and a value of thereactance element of the impedance adjustment circuit are determinedsuch that a phase-shift amount changes in accordance with a frequencyband.
 2. A phase shifter comprising: a transformer connected between afirst port and a second port and including a first coil and a secondcoil that is magnetically coupled to the first coil, the transformerincluding a parasitic inductance component; and an impedance adjustmentcircuit including a reactance element that suppresses a deviation inimpedance due to the parasitic inductance component of the transformer;wherein a coupling coefficient between the first coil and the secondcoil of the transformer and a value of the reactance element of theimpedance adjustment circuit are determined such that a phase-shiftamount in a low band is closer to 180° than 90° and that a phase-shiftamount in a high band is closer to 90° than 180°.
 3. A phase shiftercomprising: a transformer connected between a first port and a secondport and including a first coil and a second coil that is magneticallycoupled to the first coil, the transformer including a parasiticinductance component; and an impedance adjustment circuit including areactance element that suppresses a deviation in impedance due to theparasitic inductance component of the transformer; wherein a couplingcoefficient between the first coil and the second coil of thetransformer and a value of the reactance element of the impedanceadjustment circuit are determined such that a phase-shift amount in alow band is closer to 90° than 0° and that a phase-shift amount in ahigh band is closer to 0° than 90°.
 4. The phase shifter according toclaim 1, wherein the impedance adjustment circuit includes: a firstcapacitance element connected between the first port connected to thetransformer and a ground; a second capacitance element connected betweenthe second port connected to the transformer and the ground; and a thirdcapacitance element connected between the first port and the second portconnected to the transformer.
 5. The phase shifter according to claim 1,wherein the impedance adjustment circuit includes: a first capacitanceelement connected between the first port connected to the transformerand a ground; a second capacitance element connected between the secondport connected to the transformer and the ground; and a series circuitincluding a third capacitance element and an inductance element, theseries circuit being connected between the first port and the secondport connected to the transformer.
 6. The phase shifter according toclaim 4, wherein the third capacitance element is mainly defined by acapacitance between the first coil and the second coil.
 7. The phaseshifter according to claim 4, wherein the first capacitance element ismainly defined by a capacitance between wires of the first coil, and thesecond capacitance element is mainly defined by a capacitance betweenwires of the second coil.
 8. The phase shifter according to claim 1,wherein a transformer ratio between the first coil and the second coilis 1:n where n is a value other than 1; and a phase-shift amount of thephase shifter is moved toward a center of a Smith chart by a reflectioncoefficient being moved from a high-impedance side to a low-impedanceside on the Smith chart and impedance conversion by the phase shifter.9. The phase shifter according to claim 1, wherein the transformer isprovided in a single stack in which a plurality of base layers arestacked, and the first coil and the second coil are defined by conductorpatterns provided on the base layers.
 10. The phase shifter according toclaim 9, wherein the first coil and the second coil have a same or asubstantially same inside diameter, and a same or a substantially sameoutside diameter and have co-axial coil winding axes.
 11. The phaseshifter according to claim 1, further comprising a high-pass filter or alow-pass filter connected in series to the phase shifter.
 12. The phaseshifter according to claim 11, wherein the high-pass filter or thelow-pass filter includes a capacitance element and an inductanceelement, the inductance element being magnetically coupled to the firstcoil or the second coil.
 13. An impedance matching circuit comprising:the phase shifter according to claim 1; and an impedance matchingcircuit that is connected in series to the phase shifter; wherein theimpedance matching circuit performs impedance matching of an impedancewith a phase shifted by the phase shifter.
 14. The impedance matchingcircuit according to claim 13, wherein the phase shifter moves impedancein a low band to a second quadrant or a third quadrant on a Smith chart;and the impedance matching circuit moves both impedance in a high bandand the impedance in the low band toward a center of the Smith chart.15. A multi/demultiplexer comprising: the phase shifter according toclaim 1; a high-pass filter that performs high-band passing; and alow-pass filter that performs low-band passing; wherein the high-passfilter includes a shunt-connected first inductor between a signal lineand a ground and a first capacitor connected in series in a followingstage of the first inductor; the low-pass filter includes a secondinductor connected in series to a common port and a second capacitorconnected in shunt to the ground in a following stage of the secondinductor; the phase shifter is inserted between the common port and thefirst inductor; and the phase shifter shifts a phase such that thehigh-pass filter for high-band passing is substantially open in a passfrequency band of the low-pass filter for low-band passing seen from thecommon port.
 16. A multi/demultiplexer comprising: the phase shifteraccording to claim 1; and a plurality of surface acoustic wave filtersincluding a first surface acoustic wave filter and a second surfaceacoustic wave filter with mutually different pass frequency bands, thefirst surface acoustic wave filter and the second surface acoustic wavefilter each including a first port and a second port; wherein the firstport of the first surface acoustic wave filter is connected to a commonport via the phase shifter, and the second port of the first surfaceacoustic wave filter is connected to an individual port; and the phaseshifter shifts a phase such that the first surface acoustic wave filteris substantially open in a pass frequency band of the second surfaceacoustic wave filter seen from the common port.
 17. A communicationterminal apparatus comprising: a feeder circuit; and an antennaconnected to the feeder circuit; wherein between the feeder circuit andthe antenna, the phase shifter according to claim 1 is provided.
 18. Acommunication terminal apparatus comprising: a feeder circuit; and anantenna connected to the feeder circuit; wherein between the feedercircuit and the antenna, the impedance matching circuit according toclaim 13 is provided.
 19. A communication terminal apparatus comprising:a feeder circuit; and an antenna connected to the feeder circuit;wherein between the feeder circuit and the antenna, themulti/demultiplexer according to claim 15 is provided.